This invention relates to a distortion compensating apparatus and, more particularly, to a distortion compensating apparatus having a distortion compensation coefficient calculating unit, to which is input a difference signal between a reference signal that is a transmit signal and a feedback signal, for calculating a distortion compensation coefficient from an adaptive algorithm in such a manner that the difference signal will be diminished, a distortion compensation coefficient memory the stored content of which is updated by the distortion compensation coefficient calculated, and a distortion compensator for applying distortion compensation to the transmit signal based upon the distortion compensation coefficient.
In wireless communications in recent years, there is growing use of high-efficiency transmission using digital techniques. In instances where multilevel phase modulation is applied to wireless communications, a vital technique is one which can suppress non-linear distortion by linearizing the amplification characteristic of the power amplifier on the transmitting side and reduce the leakage of power between adjacent channels. Also essential is a technique which compensates for the occurrence of distortion that arises when an attempt is made to improve power efficiency by using an amplifier that exhibits poor linearity.
FIG. 27 is a block diagram illustrating an example of a transmitting apparatus in a radio according to the prior art. Here a transmit-signal generator 1 transmits a serial digital data sequence and a serial/parallel (S/P) converter 2 splits the digital data sequence alternately one bit at a time to convert the data to two sequences, namely an in-phase component signal (also referred to as an “I signal”) and a quadrature component signal (also referred to as a “Q signal”). A DA converter 3 converts the I and Q signals to respective analog baseband signals and inputs these to a quadrature modulator 4. The latter multiplies the input I and Q signals (the transmit baseband signals) by a reference carrier wave and a signal that has been phase-shifted relative to the reference carrier by 90°, respectively, and adds the results of multiplication to thereby perform quadrature modulation and output the modulated signal. A frequency converter 5 mixes the quadrature-modulated signal and a local oscillation signal to thereby effect a frequency conversion, and a transmission power amplifier 6 power-amplifies the carrier output from the frequency converter 5. The amplified signal is released into space from an antenna 7.
In mobile communications based upon W-CDMA, etc., the transmission power of the transmitting apparatus is a high 10 mW to several tens of watts, and the input/output characteristic [distortion function f(p)] of the transmission power amplifier 6 is non-linear, as indicated by the dotted line at (a) of FIG. 28. Non-linear distortion arises as a result of this non-linear characteristic, and the frequency spectrum in the vicinity of a transmission frequency f0 comes to exhibit side lobes, as indicated by the solid line in (b) of FIG. 28, leakage into adjacent channels occurs and this causes interference between adjacent channels. More specifically, owing to non-linear distortion, there is an increase in power that causes transmitted waves to leak into the adjacent frequency channels, as shown in at (b) in FIG. 28. ACPR (Adjacent Channel Power Ratio), which indicates the magnitude of leakage power, is the ratio between the power of the channel of interest, which is the area of the spectrum between the one-dot chain lines A and A′ at (b) in FIG. 28, and the adjacent leakage power, which is the area of the spectrum between the two-dot chain lines B and B′, that leaks into the adjacent channel. Such leakage power constitutes noise in other channels and degrades the quality of communication of these channels. Such leakage must be limited to the utmost degree.
Leakage power is small in the linear region [see (a) in FIG. 28] of a power amplifier and large in the non-linear region. Accordingly, it is necessary to broaden the linear region in order to obtain a transmission power amplifier having a high output. However, this necessitates an amplifier having a performance higher than that actually needed and therefore is inconvenient in terms of cost and apparatus size. Accordingly, a radio apparatus that has come to be adopted is equipped with a distortion compensating function that compensates for distortion of the transmission power.
FIG. 29 is a block diagram of a transmitting apparatus having a digital non-linear distortion compensating function that employs a DSP (Digital Signal Processor). Here a group of digital data (a transmit signal) sent from the transmit-signal generator 1 is converted to two signal sequences, namely I and Q signals, by the S/P converter 2, and these signals enter a distortion compensator 8 constituted by a DSP. The distortion compensator 8 has a distortion compensation coefficient memory 8a for storing distortion compensation coefficients h(pi) (i=0˜1023) conforming to power levels pi of a transmit signal x(t); a predistortion unit 8b for subjecting the transmit signal to distortion compensation processing (predistortion) using a distortion compensation coefficient h(pi) that is in conformity with the level of the transmit signal; and a distortion compensation coefficient calculation unit 8c for comparing the transmit signal x(t) with a demodulated signal (feedback signal) y(t), which has been obtained by demodulation in a quadrature detector described later, and for calculating and updating the distortion compensation coefficient h(pi) in such a manner that the difference between the compared signals will approach zero.
The signal that has been subjected to predistortion processing in the distortion compensator 8 is input to the DA converter 3. The latter converts the input I and Q signals to analog baseband signals and applies the baseband signals to the quadrature modulator 4. The latter multiplies the input I and Q signals by a reference carrier wave and a signal that has been phase-shifted relative to the reference carrier by 90°, respectively, and sums the results of multiplication to thereby perform quadrature modulation and output the modulated signal. The frequency converter 5 mixes the quadrature-modulated signal and a local oscillation signal to thereby effect a frequency conversion, and the transmission power amplifier 6 power-amplifies the carrier signal output from the frequency converter 5. The amplified signal is released into the atmosphere from the antenna 7.
Part of the transmit signal is input to a frequency converter 10 via a directional coupler 9, whereby the signal undergoes a frequency conversion and is input to a quadrature detector 11. The latter performs quadrature detection by multiplying the input signal by a reference carrier wave and a signal that has been phase-shifted relative to the reference carrier by 90°, reproduces the I, Q signals of the baseband on the transmitting side and applies these signals to an AD converter 12. The latter converts the applied I and Q signals to digital data and inputs the digital data to the distortion compensator 8. By way of adaptive signal processing using the LMS (Least Mean Square) algorithm, the distortion compensator 8 compares the transmit signal before distortion compensation with the feedback signal demodulated by the quadrature detector 11 and proceeds to calculate and update the distortion compensation coefficient h(pi) in such a manner that the difference between the compared signals will become zero. By thenceforth repeating this operation, non-linear distortion of the transmission power amplifier 6 is suppressed to reduce the leakage of power between adjacent channels.
FIG. 30 is a diagram useful in describing distortion compensation processing by an adaptive LMS. Reference numeral 15a denotes a multiplier (which corresponds to the predistortion unit 8b in FIG. 29) for multiplying the transmit signal x(t) by a distortion compensation coefficient hn−1 (p). Reference numeral 15b represents a transmission power amplifier having a distortion function f(p). Reference numeral 15c denotes a feedback loop for sending back the output signal y(t) from the transmission power amplifier. Reference numeral 15d denotes an arithmetic unit (amplitude-to-power converter) for calculating the power p [=x(t)2] of the transmit signal x(t). Reference numeral 15e represents a distortion compensation coefficient memory (which corresponds to the distortion compensation coefficient memory 8a of FIG. 29) 15e for storing the distortion compensation coefficients that conform to the power levels of the transmit signal x(t). The memory 15e outputs the distortion compensation coefficient hn−1(p) conforming to the power p of the transmit signal x(t) and updates the distortion compensation coefficient hn−1(p) by a distortion compensation coefficient hn(p) found by the LMS algorithm.
Reference numeral 15f denotes a complex-conjugate signal output unit, 15g a subtractor that outputs the difference e(t) between the transmit signal x(t) and the feedback demodulated signal y(t), 15h a multiplier that performs multiplication between e(t) and u*(t), 15i a multiplier that performs multiplication between hn−1(p) and y*(t), 15j a multiplier that performs multiplication by a step-size parameter μ, and 15k an adder that adds hn−1(p) and μe(t)u*(t). Reference numerals 15m, 15n, 15p denote delay units. A delay time D, which is equivalent to the length of time from the moment the transmit signal x(t) enters to the moment the feedback (demodulated) signal y(t) is input to the subtractor 15g, is added onto the input signal.
Reference numerals 15f and 15h˜15j construct a rotation calculation unit 16. A signal that has sustained distortion is indicated at u(t). The delay time D set in the delay units 15m, 15n, 15p is decided so as to satisfy D=D0+D1, where D0 represents the delay time in the transmission power amplifier 15b and D1 the delay time of the feedback loop 15c. If the delay time D cannot be set correctly, the distortion compensating function will not operate effectively. Further, the larger the setting error of the delay time, the larger the sidelobes and the greater the leakage power to adjacent channels.
The arithmetic operations performed by the arrangement set forth above are as follows:hn(p)=hn−1(p)+μe(t)u*(t)e(t)=x(t)−y(t)y(t)=hn−1(p)x(t)f(p)u(t)=x(t)f(p)=h*n−1(p)y(t)P=|x(t)|2 where x, y, f, h, u, e represent complex numbers and * signifies a complex conjugate. By executing the processing set forth above, the distortion compensation coefficient h(p) is updated so as to minimize the difference e(t) between the transmit signal x(t) and the feedback demodulated signal y(t), and the coefficient eventually converges to the optimum distortion compensation coefficient so that compensation is made for the distortion in the transmission power amplifier. FIG. 31 is a diagram showing the overall structure of a transmitting apparatus expressed by x(t)=I(t)+jQ(t). Components in FIG. 31 identical with those shown in FIGS. 29 and 30 are designated by like reference characters.
As mentioned above, the principle of a distortion compensating apparatus is to feed back and detect a carrier obtained by quadrature modulation of a transmit signal, digitally convert and compare the amplitudes of the transmit signal and feedback signal, and update the distortion compensation coefficient in real time based upon the result of the comparison. In accordance with this method of non-linear distortion compensation, distortion can be reduced. As a result, leakage power can be kept low even with operation at high output and in a non-linear region, and it is possible to improve power load efficiency.
Even if the delay time D is set correctly so as to satisfy D=D0+D1, there are instances where a favorable, stabilized distortion compensation operation cannot be obtained and unnecessary out-of-band power is produced. The cause is clock jitter produced by thermal noise and other disturbances in the analog system that includes the AD and DA converters. When clock jitter occurs, the phase of the feedback signal fluctuates violently and this has an influence upon convergence of distortion compensation coefficient.
Owing to jitter, clock speed changes repeatedly, sometimes attaining a high speed and sometimes a low speed. Consequently, the phase difference of the feedback signal relative to the reference signal varies, as illustrated in FIG. 32 by way of example. With the conventional distortion compensating apparatus, phase fluctuation due to clock jitter is not taken into consideration. As a result, the distortion compensation coefficient undergoes unstable fluctuation within the range of phase fluctuation. Since the distortion compensation coefficient is multiplied by the transmit signal, this brings about the generation of unwanted waves.